System and method for improving polarization matching on a cellular communication forward link

ABSTRACT

The present invention provides a method of controlling a plurality of beam patterns radiated by a base station in a wireless communication system. The method includes receiving at least one signal from a mobile station at the base station, determining estimated attributes of the at least one signal received by the base station. Smoothed versions of the estimated attributes are calculated in accordance with a predetermined set of criteria are calculated. A set of weighted signal parameters are generated to describe a polarization state of the at least one signal received from the mobile station. The generated weighed signal parameters are applied to a signal transmitted by the base station such that the transmitted signal substantially matches the polarization state of the at least one signal from the mobile station.

RELATED APPLICATION DATA

[0001] The present application is related to the disclosures provided inU.S. Provisional application No. 60/202,197 entitled SYSTEM AND METHODFOR PROVIDING POLARIZATION MATCHING ON A CELLULAR COMMUNICATION FORWARDLINK filed May 5, 2000; U.S. Provisional application No. 60/225,388entitled SYSTEM AND METHOD FOR IMPLEMENTING POLARIZATION MATCHING ON ACELLULAR COMMUNICATION FORWARD LINK filed Aug. 15, 2000; and U.S.Provisional application No. 60/268,700 entitled “SYSTEM AND METHOD FORIMPLEMENTING POLARIZATION MATCHING FOR CELLULAR COMMUNICATIONS SYSTEMSHAVING MULTIPLE ANTENNA ELEMENTS”filed on Feb. 15, 2001. The contents ofthese applications are hereby expressly incorporated by reference intheir entireties.

BACKGROUND RESERVATION OF COPYRIGHT

[0002] The disclosure of this patent document contains material which issubject to copyright protection. The copyright owner has no objection tothe facsimile reproduction by anyone of the patent document or thepatent disclosure, but otherwise reserves all copyrights whatsoever.

FIELD OF THE INVENTION

[0003] This present invention, in certain respects, relates to the fieldof cellular communications. In other respects, the present inventionrelates to polarization matching: on forward link transmissions of acellular communication system and methods to implement the polarizationmatching for a cellular communication system.

DESCRIPTION OF BACKGROUND INFORMATION

[0004] Today's cellular communication systems are subjected toever-increasing user demands. Current subscribers are demanding moreservices and better quality while system capacities are being pushed totheir limits. In striving to achieve higher capacities and better gradesof service, it becomes necessary to optimize transmission integrity bydecreasing transmission losses wherever possible.

[0005] Typically, for each geographic cell, cellular communicationsystems employ a base station (BS) with an omni-directional antenna thatprovides signal coverage throughout the cell. An alternative approachangularly divides the geographic cells into sectors (i.e., sectoring)and deploys BS antennae that radiate highly-directive narrow beampatterns to cover designated sectors. The directive beam patterns can benarrow in both the azimuthal and elevation plane and, by virtue of theirdirectional gain, enable mobile stations (MSs) to communicate with theBS at longer distances.

[0006] The narrow beams used to form beam patterns for given coverageareas are optimized to improve performance of the wireless network.Optimization may include the polarization of the beams to enhanceperformance. Traditionally, BSs employ a plurality ofvertically-polarized antennae, which complement the vertically-polarizedantennae of most mounted MSs (e.g., non-handheld MSs). Thisconfiguration, however, is insufficient in accommodating signals fromhand-held MSs, as well as cars with non-vertical antennas.

[0007] Generally, hand-held MSs have transmit antennae that are linearlypolarized. The polarization is randomly distributed depending upon theposition in which the MS is held. For example, the polarization ofhand-held MS antenna may vary up to 20° degrees from the horizontal tothe vertical plane relative to the BS antennae. Thus, there exists anpolarization mismatch between hand-held MSs and BS antennae. Such apolarization mismatch can cause an average transmission loss of up to 7dB and instantaneous transmission losses of up to 9 dB.

[0008] In an effort to overcome such costly transmission losses, someBSs incorporate multiple polarization-diverse antennae with combinercircuitry to “match” the polarization of the MS-to-BS transmissions(i.e., reverse link transmissions). For example, FIG. 1 illustrates areceive portion of a BS antenna system 100 employingpolarization-diverse receive antenna elements 102, 104. The receiveelements 102, 104 are configured to accommodate two opposing (i.e.,orthogonal) linearly-slanted polarized states (i.e., ±45° linearpolarization). The signals received by both types of antenna elements102, 104 are applied to a diversity combining circuit 106, whichdetermines a maximum value in a preferred signal parameter (e.g.,signal-to-noise ratio, SNR) between the two signals received fromantenna elements 102, 104. In this manner, the BS is able to limittransmission losses due to polarization mismatches by matching thereverse link transmission to one of two possible polarization states(i.e., ±45°).

[0009] Because hand-held MSs are not generally equipped with multipleantennae, such polarization matching schemes cannot be implemented inMSs to compensate for polarization mismatches during BS-to-MStransmissions (i.e., forward link transmissions). This results in costlytransmission losses in the forward link. What is needed, therefore, iseffective polarization matching on forward link transmissions tomitigate transmission losses due to the polarization mismatch betweenhand-held MSs and BSs.

BRIEF DESCRIPTION OF THE DRAWINGS

[0010] The above and other objects, features and advantages of thepresent invention are further described in the detailed descriptionwhich follows, with reference to the drawings by way of non-limitingexemplary embodiments of the present invention, wherein like referencenumerals represent similar parts of the present invention throughout theseveral views and wherein:

[0011]FIG. 1 is a high level diagram depicting a first type of antennaarrangement;

[0012]FIG. 2B depicts different antenna polarization arrangement units;

[0013]FIG. 3A depicts an active radiator unit;

[0014]FIG. 3B depicts another antenna arrangement;

[0015]FIG. 4A is a high level diagram illustrating a BS antenna systemcapable of shaping composite beams;

[0016]FIG. 4B is a diagram of a composite beam;

[0017]FIGS. 5A, 5B, 5C, 5D are high level diagrams illustratingembodiments of the present invention;

[0018]FIG. 6 is a high level diagram depicting the polarization matchingprocessor;

[0019]FIG. 7 is a flowchart of one embodiment of the polarizationmatching processor algorithm;

[0020]FIG. 8 is a high level diagram depicting a trial system;

[0021] FIGS. 9-12 depict graphs of a plurality of trial tests;

[0022]FIG. 13 illustrates a received vector representing a RF signalpolarized in a direction θ;

[0023]FIG. 14 is a graph that illustrates the dependence of ABS (Σ) andABS (Δ) versus β;

[0024]FIG. 15 is a graph of the log ratio Σ/Δ;

[0025]FIG. 16 is a high level diagram depicting an embodiment of βestimation;

[0026]FIG. 17 is a high level diagram depicting another embodiment of βestimation;

[0027]FIG. 18 is a high level diagram depicting an embodiment of a TDMAsystem (single carrier);

[0028]FIG. 19 is a high level diagram depicting polarization matchingmechanism;

[0029]FIG. 20 is a high level diagram illustrating an embodiment of theinvention;

[0030]FIG. 21 is a high level diagram illustrating an embodiment of theinvention;

[0031]FIG. 22 is a high level diagram illustrating an embodiment of theinvention;

[0032]FIG. 23 is a phasor diagram illustrating the principle of equalpower realization;

[0033]FIG. 24 compares the loading of power amplifiers;

[0034] FIGS. 25A-25C illustrate the histograms;

[0035]FIG. 26 illustrates the simulation results on diversity gains;

[0036]FIG. 27 is a high level diagram illustrating an embodiment of theinvention;

[0037]FIG. 28 is a high level diagram illustrating an embodiment of theinvention;

[0038]FIG. 29 is a high level diagram illustrating an embodiment of theinvention;

[0039]FIG. 30 illustrates the behavior of diversity gain;

[0040]FIG. 31 is a high level diagram illustrating an embodiment of theinvention; and

[0041]FIG. 32 is a high level diagram illustrating an embodiment of theinvention.

DETAILED DESCRIPTION

[0042] The present invention, for example, utilizes the informationmeasured and processed on the receive (Rx) polarization to estimate theaverage required transmit (Tx) polarization. The average polarizationstate of the waves is not frequency dependent, and can be inferred fromthe reverse link, in order to determine weights in the forward linktransmission. The Polarization matching method for Base TranceiverSystem (BTS) forward link transmission management improves the powervalue and efficiency along with reduced interference. Additionalimportant benefits stem from the improved soft handoff (SHO) zoneperformance.

[0043]FIG. 2 illustrates a representative embodiment of an activeantenna array configuration for a BS, which is described in “ACTIVEANTENNA ARRAY CONFIGURATION AND CONTROL FOR CELLULAR COMMUNICATIONSYSTEMS,” application Ser. No. 09/357,844 filed on Jul. 21, 1999. Asdepicted in FIG. 2, antenna array 250 comprises a combination of twoactive transmit antenna elements 255A, 255B and two active receiveantenna elements 260A, 260B arranged in a single vertical (columnar)array. The two active transmit antenna elements 255A, 255B and twoactive receive antenna elements 260A, 260B are preferably printedelemental radiators having a multi-layer configuration and sealed by anepoxy-fiberglass radome.

[0044] By incorporating separate transmit antenna elements 255A, 255Band receive antenna elements 260A, 260B within a single array, the BS iscapable of achieving full transmission and reception functionality forcellular operations while eliminating the need for independenttransmission and reception antenna arrays. In doing so, antenna array250 achieves full BS functionality in a streamlined and compact design.

[0045] The spatial separation of the transmit 255A, 255B and receiveantenna elements 260A, 260B within the array also avoids theintermodulation interference on the receive portion caused by the highpower transmit signals, as stated above with respect to conventionalcombined-element systems. The spatial separation also providesflexibility in BS transmission and reception optimization schemes, suchas, for example, independent gain control and beam-shaping, which islimited in combined-element systems. In addition, the separation alsoobviates the need for signal discriminating hardware, such as duplexersand complex transmit and receive filters which, in attempting to isolateand filter the respective signals from combined transmit/receive antennaelements, operate in a relatively lossy and inefficient manner. Suchspatial separation also results in additional isolation between thereceive and transmit signals.

[0046]FIG. 2 further illustrates that, within the vertical arrangement,the antenna elements are disposed in an alternating fashion such that afirst transmit antenna element 255A is followed by a first receiveantenna element 260A and a second transmit antenna element 255B isfollowed by a second receive antenna element 260B. The interleaving ofthe transmit 255A, 255B and receive 260A, 260B antenna elements withinthe array enables the optimal vertical separation distance S to beestablished. Optimal vertical separation distance S is the verticaldistance between like antenna elements which, for a given frequency,maximizes the main lobe gain of a signal while minimizing thecontribution of minor lobes. The optimal vertical separation distance Scan vary. For example, in a personal communication system (PCS), S maybe from 0.70λ to 0.95λ.

[0047] Additionally, the transmit 255A, 255B and receive 260A, 260Bantenna elements within the array antenna are configured to producepolarized radiated patterns. Polarization of a radiated pattern in aspecified direction results in the maximum gain of the pattern along thespecified direction.

[0048] Because of multipath considerations, coupled with the relativelylow transmit power of MSs, antenna array 250 provides the additionalbenefit that it is configured to provide polarization diversity for boththe BS transmit antenna elements 255A, 255B and the BS receive antennaelements 260A, 260B. Specifically, each of transmit antenna elements255A, 255B and receive antenna elements 260A, 260B comprise a pair oforthogonally polarized antenna elements. Polarization diversitytypically requires two antenna elements that are orthogonally polarized.The effectiveness of polarization diversity depends on the similarity ofthe radiated patterns transmitted or received by the two antennaelements and on the equality of the average signal level transmitted orreceived by the elements. For example, as stated above, hand-held MSspossess antennae that are linearly polarized with a random distributiondepending upon the position in which the MS is held. As such, antennaarray 250 exploits these polarization states by configuring each of theBS transmit antenna elements 255A, 255B and each of the receive antennaelements 260A, 260B to accommodate two opposing (i.e., orthogonal)linearly-slanted polarized states (i.e., ±45° linear polarization).

[0049] It is to be understood that the specific arrangement of antennaarray 250 may be modified to provide redundancy or otherwise enhance theattributes and characteristics of the array configuration. For example,antenna array 250 may be augmented by stacking combinations of the arrayto achieve antenna elements arranged in an 8×1, 12×1, or 16×1 arrayconfiguration. This arrangement, therefore, provides a compactsingle-column array antenna configuration for cellular communicationshaving full transmission and reception capabilities. Further, thearrangement enables independent transmit and receive gain control andbeam-shaping, minimizes transmit intermodulation interference, andprovides both transmit and receive polarization diversity.

[0050]FIG. 3A depicts a representative embodiment of an Active RadiatingUnit (ARU) 300, which is described in the commonly-assigned applicationentitled “SCALABLE CELLULAR COMMUNICATIONS SYSTEM,” application Ser. No.09/357,844 filed on Jul. 21, 1999. The ARU 300 comprises a modularantenna unit having a transmit path and a receive path. The transmitpath incorporates a power amplifier (PA) 302, which is electricallycoupled to a transmit band-pass filter 304. The transmit filter 304 is,in turn, electrically coupled to a transmit antenna 306. The transmitantenna 306 may be configured for a variety of operations, including,for example, vertical or dual slanted-linear polarization, as indicatedabove in antenna arrays 100, 250. Similarly, the receive path implementsa receive antenna 316, which is electrically coupled to a receivebandpass filter 314. The receive antenna 316 may also be configured fora variety of operations, including, for example, vertical or dualslanted-linear polarization, as indicated above in antenna arrays 100,250. The receive bandpass filter 314 is subsequently coupled to alow-noise amplifier (LNA) 312. The ARU 300 may also include monitoringand control sub-units as well as power conditioning sub-units in orderto provide supervisory control, management functionality, and optimalperformance. As such, the ARU 300 provides transmission and receptionpath portions within a single modular unit.

[0051]FIG. 3B illustrates antenna array 350, deploying a plurality ofARUs 300 in an 8×1 (columnar) arrangement. Inputs to the array 350 arefacilitated by two corporate feeds, which respectively interconnect allthe transmit antenna elements and all the receive antenna elements. Asstated above with respect to ARU 300, the transmit elements may bevertically polarized and the receive antenna elements may belinearly-slant polarized (i.e., ±45° linear polarization).Alternatively, antenna array 350 may be configured to have transmit andreceive polarization diversity by configuring both the transmit antennaelements and the receive antenna elements to have linearly-slantpolarization.

[0052]FIG. 4A depicts a representative embodiment of a BS antenna system400 comprising a multi-columnar antenna arrangement 410 coupled to atransform matrix 420, as described in the commonly-assigned applicationentitled “ACTIVE ANTENNA ARRAY CONFIGURATION AND CONTROL FOR CELLULARCOMMUNICATION SYSTEMS,” filed on Jul. 21, 1999. As illustrated in FIG.4A, the antenna arrangement 410 and the transform matrix 420 areconfigured for either transmit or receive BS operations. The transformmatrix 420 comprises a plurality of beam ports 411 on a beam-plane side412 of the matrix 420 and a plurality of antenna ports 413 on theantenna-plane side 414 of the matrix 420. Each column array (column 1,2, 3, and 4) of the multi-columnar array arrangement 410 is coupled tothe matrix 420 through the antenna ports 413. During BS system 400transmission, this configuration enables the transform matrix 420 toreceive signals, which include relative amplitude and phase information,from the beam-plane side 412. Based on this information, the matrix 420transforms the beam-plane signals into signals appropriate for theradiating antenna elements (column 1, 2, 3, and 4) and delivers suchsignals to all the antenna ports 413. The antenna elements (column 1, 2,3, and 4) within the columns of multi-column array 410, then radiatenarrow shaped beam patterns in different directions in accordance withthe transformed signals. Conversely, during BS system 400 reception, thetransform matrix 420 receives signals from the antenna-plane 414 ports413 and transforms them into signals appropriate for processing. Assuch, the illustrated system 400 forms a plurality of narrow beampatterns that span different angular directions for a given axis.

[0053]FIG. 4A further illustrates that each of the beam ports 411 on thebeam-plane side 412 of the transform matrix 420 is coupled to anamplitude or gain adjusting element 430 and a phase adjusting element440. Elements 430, 440 allow for amplitude/gain and phase adjustments tobe made in order to control the shape of the antenna beam patterns, asindicated in FIG. 4B.

[0054]FIG. 4B illustrates a typical composite beam pattern radiated by aBS system, such as BS antenna system 400 depicted in FIG. 4A. Thetransform matrix 420 supplies signals to the antenna ports 413, whichenables the antenna elements (column 1, 2, 3, and 4) to form fourindividual beams. The aggregate effect of these individual beams is theenvelope composite beam, as indicated in FIG. 4B. As stated above,amplitude/gain adjusting elements 430 and phase adjusting elements 440make it possible to control the shape of the antenna beam patterns. Theamplitude level at which the individual beams (i.e., sub-beams)intersect is called the cross-over level. The position of the cross-overlevel depends, at least in part, on the optimal separation distance ofthe antenna elements contained in the antenna array 410.

[0055]FIG. 5A illustrates an embodiment of the present invention,incorporating some of the features noted above to provide polarizationmatching on a forward link transmission of a cellular communicationsystem. In particular, FIG. 5A depicts a BS antenna system 500comprising a multi-columnar antenna arrangement 510 coupled to transmittransform matrices 520, 522 and receive transform matrices 524, 526,respectively. The multi-columnar antenna arrangement 510 includes one ormore transmit-receive antenna sets; each set includes two transmitantenna elements and two receive antenna elements. The illustratedarrangement is used to achieve polarization matching on both, thetransmit and receive portion of the system 500. For example, asindicated in FIG. 5A both transmit and receive antenna elements comprisetwo opposing (i.e., orthogonal) linearly-slanted polarized states (i.e.,±45° linear polarization). It is to be noted that other antennaarrangements along with different polarization orientations may beprovided in order to achieve diversity on the transmit and receiveportions. For example, FIG. 5A may be altered to illustrate anembodiment of a single column, without transform matrices, and withweights 530, 532, 534, 536 comprising a single amplitude/gain and phaseadjusters, such as attenuator and phase shifters, and further comprisinga diversity combiner and a splitter.

[0056] Each of the transmit and receive antenna elements within thecolumns of the array arrangement 510 are associated with the antennaports corresponding to the respective transmit 520, 522 and receive 524,526 matrices. The antenna ports are coupled to a host of amplitude/gainand phase adjustments 512, 514, 516, 518. Conversely, the beam ports ofthe transmit 520, 522 and receive 524, 526 matrices are coupled to aseparate host of amplitude/gain and phase adjustments 530, 532, 534,536. As such, the polarization of each radiated beam pattern can becontrolled separately. Alternatively, the configuration may omit theamplitude/gain and phase adjustments, and thus not control thepolarization of the radiated beam patterns.

[0057]FIG. 5A further illustrates an adaptive measurement and controlportion 540 coupled to antenna port amplitude/gain and phase adjustments512, 514, 516, 518 and to beam ports 530, 532, 534 and 536. The adaptivemeasurement and control portion 540 may be provided with a “fast”mechanism for quickly adapting to fading signals on the received reverselink by adjusting the receive amplitude/gain and the phase adjustments516, 518 at the antenna plane. The adaptive measurement and controlportion 540 further comprises a “slow” mechanism that follows thephysical movements of a particular MS and averages the fading of thereceived reverse link. As indicated in FIG. 5A, adaptive measurement andcontrol portion 540 is also applied to amplitude/gain and phaseadjustments 512, 514 on the transmit portion of system 500 to vary thepolarization on the forward link in an effort to match the short-termaverage polarization of the MS.

[0058] Specifically, as depicted in system 550 of FIG. 5B, the signaldetected from the MS on each of the receive antenna elements 507 isamplitude/gain adjusted and/or phase adjusted by amounts a1 and a2,which may be equal to each other. The adjusted signal for each of thereceive antenna elements is combined and adaptively controlled by themeasurement and control portion 540, which drives the values of a1 anda2. This diversity combining may be applied at baseband, which requirestwo full receiver chains, at RF, or at IF.

[0059] The amplitude/gain adjustments and/or phase adjustments driven bymeasurement and control portion 540 are subsequently supplied totransform 560. The transform 560 first transforms amplitude/gainadjustments and/or phase adjustments in accordance with the differencesin gain of the transmitting antenna elements. A low-pass filter portion565, coupled to transform 560, then averages the fast control variationsand responds only to the slow variations resulting from the physicalattitude changes of the MS. The output of low-pass filter portion 565generates transmit amplitude/gain adjustments and/or phase adjustmentsb1 and b2, which are applied to each MS at baseband.

[0060] It is to be understood that for single RF transmissions (e.g.,single RF TDMA channel), the transmission may be adjusted in the RF sideof the BS or by controlling the associated amplifiers in the chain (see,for example, FIG. 3A). It is also understood that for multi-channeltransmissions, the signal may be split into two branches, adjusted forpolarization matching, and combined with the associated transmissionchain.

[0061] Thus, this embodiment utilizes information driven by the receivedreturn link signals to ensure that the transmitted signal polarizationmatches the polarization of the incoming signal. In doing so, forwardlink operation is enhanced and transmission losses due to polarizationmismatches are significantly reduced.

[0062] The base station antenna may be adapted to be used with variouscommunications technologies, such as GSM and CDMA. To illustrate, FIG.5C is another embodiment of the present invention, incorporating some ofthe features noted above to provide polarization matching on a forwardlink transmission of a cellular communication system, such as GSM. Inparticular, this embodiment adjusts the amplitude/gain and/or phase ofthe forward link transmission based on the polarization of the reverselink transmission. This embodiment is applied per MS user (i.e.,time-slot), and per carrier, which may be applied at baseband, IF, orRF. This exemplary embodiment relates to RF.

[0063] In particular, FIG. SC depicts a BS antenna system 5000comprising a multi-columnar antenna arrangement 5050. The output powerof the antenna arrangement 5050 is split into two orthogonalpolarizations, such as, for example, opposite linearly-slanted polarizedantenna elements (i.e., ±45° linear polarizations). The power istransmitted according to the originally received polarization. Theillustrated arrangement is used to achieve polarization matching on boththe transmit and receive portion of the system 5000. Other antennaarrangements along with different polarization orientations may also beprovided.

[0064] The signals from the receive portion of the antenna arrangement5050 are routed to amplitude/gain adjustments and/or phase adjustments5040, which may be used to control the carriers. The receiveamplitude/gain adjustments and/or phase adjustments 5040 areelectrically coupled with splitters 5045, which replicate their inputsignal. For instance, in FIG. 5C, each of the splitters 5045 replicatesthe input signal four times, and thus outputs four similar signals.These output signals are transmitted to transmit and receive controller5000 via down converter 5010 and switches 5005.

[0065] A processing unit may comprise splitters 5020, amplitude/gainadjustments and/or phase adjustments 5025, combiners 5035, and processor5030. The splitters 5020 receive signals, as input, that are transferredfrom the controller 5000 via the switches 5005 and up converters 5015.The splitters 5020 replicate their input signals twice, and thus each ofthe splitters 5020 output two similar signals. These output signals areweighted with the amplitude/gain adjustments and/or phase adjustments5025. The processor 5030 determines the weights (per user per carrier)based, in part, on input parameters, such as channel status, diversity,equalizer, dominant coefficients, relative levels, and maximal ratiocombining complex coefficients. That is, the down-converter 5010demodulates each of the (4) received carrier pairs (received each viatwo orthogonally polarized antennas. The down converter 5010 alsooptimally combines each carrier pair obtaining Rx diversity. This isrepeated sequentially per time slot. Thus, each pair of received signalsresults in one output of baseband information, per carrier. Furthermore,the down converter 5010 performs equalizer functions. As such, downconverter 5010 estimates channel parameters. The channel parameters arethen passed to the processor 5030, which processes the information, peruser, to estimate channel dynamics. Also, the receiver optimal combiningweights are conveyed to the processor 5030. The processor 5030 thenfilters the information to produce the Transmit polarization matchingweights, which configure the amplitude/gain adjustments and/or phaseadjustments 5025.

[0066] Combiners 5035 then sum up the weighted carriers in order for thecarriers (i.e., RF signals) to be amplified for each orthogonalpolarized antenna. Thus, the output of the combiners 5035 resembles theform of one of the following equations:

RF ₁=Σ(a _(k))(RF _(k));

[0067] or

RF ₂=Σ(b _(k))(RF _(k)),

[0068] wherein a and b are the weights, and RF_(K) are the outputs of upconverters 5015. It should be noted that processor 5030 applies theweights a_(k) and b_(k) at the mid-point of time frame slots. Theprocessor 5030 receives timing information from the parameters, as notedabove, to determine the time frame slots, and the mid-point of the timeframe slots. In particular, it may be assumed that the slot timing isthe mid-point between slots, or that clocked timing logic circuitrycalculates the time-shift needed to determine the mid-point, andproduces a mid-point-pulse. The timing requirements may be standardspecific, such as for GSM.

[0069] The outputs of the combiners 5035 are then transferred totransmit amplitude/gain adjustments and/or phase adjustments 5040.Finally, the outputs of the transmit amplitude/gain adjustments and/orphase adjustments 5040 are transferred to the cross-polarized antennaarrangement 5050. The output transmission, however, may need to beramped-up or ramped-down because of the weights, which vary according tothe MS users. This ramping may be applied using the amplification/gainand phase adjustments 5025, and allows for smoothed transitions betweenstates rather than abrupt switching.

[0070] Thus, this embodiment utilizes information driven by the receivedreturn link signals to ensure that the transmitted signal polarizationmatches the polarization of the incoming signal. In doing so, forwardlink operation is enhanced and transmission losses due to polarizationmismatches are significantly reduced.

[0071]FIG. 5D is another embodiment of the present invention,incorporating some of the features noted above to provide polarizationmatching on a forward link transmission of a cellular communicationsystem, such as CDMA. In particular, this embodiment adjusts theamplitude/gain and/or phase of the forward link transmission based onthe polarization of the reverse link transmission. This embodiment isapplied per MS user (i.e., CDMA channel element) in baseband. That is,CDMA transmissions are overlapping in the frequency domain (even for asingle carrier). The separate user coded transmissions are generated inbaseband, then summed and the aggregate is up-converted and transmittedin RF. Thus, operation in RF affects all users, and cannot be realizedon a per-user basis. The per-user operation may be applied prior to thesummation point, i.e., at baseband.

[0072] In a TDMA system (i.e., GSM) the application in RF is possiblefor single carrier transmissions, since the optimal polarizationmatching is performed sequentially per user in consecutive time-slots.With two or more GSM carriers, then the sequential RF polarizationmatching may be done prior to the RF summation node of the singlecarriers.

[0073]FIG. 18 depicts an embodiment of a single carrier as used with aTDMA system, such as GSM or IS-54. The system may receive the timinginformation per user (i.e., location of the TDMA time slot within theTDMA frame). A narrow band tuned receiver, which is used as alogarithmic detector, provides the signals measurements. The processorcontrols the switch timing and computes the polarization vector(sign—magnitude) estimate based, in part, on Σ-Δ and RF signal levelmeasurements.

[0074]FIG. 5D depicts a BS antenna system 5100 comprising amulti-columnar antenna arrangement 5140. The output power of the antennaarrangement 5140 is split into two orthogonal polarizations, such as,for example, opposite linearly-slanted polarized antenna elements (i.e.,±45° linear polarizations). The power is transmitted according to theoriginally received polarization. The illustrated arrangement is used toachieve polarization matching on both the transmit and receive portionsof the system 5100. Other antenna arrangements along with differentpolarization orientations may also be provided.

[0075] The signals from the receive portion of the antenna arrangement5140 are routed to amplitude/gain adjustments and/or phase adjustments5130, which may be used to control the carriers. The receiveamplitude/gain adjustments and/or phase adjustments 5130 areelectrically coupled with down converters 5135 that are electricallycoupled with CDMA channel elements 5100. In this embodiment, the numberof users comprise “n.” The channel elements 5100 output modulated I andQ components per user. That is, the channel elements 5100 are MODEMs(one per user transmissions) that both transmit and receive. RAKEreceivers within the channel elements 5100 comprise channel stateparameters, which reside in the fingers searched and weighted, per user.The Rx and Tx are parallel processes; thus, Rx1 and Rx2 are used todemodulate the reverse link signals. The I and Q components representschematically the Tx quadrature components per user.

[0076] The weighing of the polarization matching may be applieddigitally at baseband, and then summed by combiners 5115. Returning toFIG. 5D, a processing unit may comprise amplitude/gain and/or phaseadjustments 5105, combiners 5115, and processor 5110. The I and Qcomponents are then duplicated and each component coupled with anamplitude/gain and/or phase adjustment 5105. In other words, themodulated I and Q components for each user are multiplied by weights(i.e., polarization coefficients) a_(k) and b_(k). The processor 5110,which may be an ASIC or a DSP device, determines the weights of theamplitude/gain and/or phase adjustments 5105, as noted above, based, inpart, on input parameters (per channel element). The input parametersmay include the number of fingers, and their relative amplitudes andphases (this, for example, is embedded in the RAKE complex weights).From these parameters, the processor 5110 determines the channeldynamics. For example, the rate of variations of the complex weightsrelate to the user speed; and the number of dominant fingers and thevariations in the level of the fingers determine the type of channel.Combiners 5115 then sum up the weighted carriers into two complexbaseband outputs, such as: Baseband₁=Σ(a_(n) )(I _(n)+jQ_(n)); andBaseband₂=Σ(b_(n))(I_(n)+jQ_(n)), wherein a and b are the weights, and Iand Q are the outputs of the channel elements 5100.

[0077] The outputs of the combiners 5115 are transmitted to modulators5120, which accept the weighted sum of I and Q components. In turn, themodulators 5120 are electrically coupled to up-converters 5125.Up-converters 5125 are electrically coupled to transmit amplitude/gainadjustments and/or phase adjustments 5130, which are used for poweramplification. Finally, the outputs of the modulators 5120 aretransmitted to the cross-polarized transmit antenna arrangement 5140(i.e., two antennas) via the electrically coupled connection.

[0078] The embodiments illustrated in FIGS. 5(A)-5(D) utilizeinformation driven by the received return link signals to ensure thatthe transmitted signal polarization matches the polarization of theincoming signal. In doing so, forward link operation is enhanced andtransmission losses due to polarization mismatches are significantlyreduced.

[0079] The processor of FIGS. 5C and 5D, 5030 and 5110, respectively,detailed herein produces the polarization matching controls of theforward link, which serve for weighting the polarized transmissions.This processor is applicable to CDMA, GSM and other multiple accessregimes. The polarization matching processors 5030 and 5110, forexample, estimates the best instantaneous transmit polarization, peruser, at the BTS. Since the transmitted polarization is linear, thepolarization matching processor produces the orientation of thepolarization, per user.

[0080] In an FDM Cellular/PCS system, the polarization matchingprocessor operates on the dual (orthogonal) polarization reverse linkreceptions, to produce forward link polarization matching weights, peruser. FIG. 6 illustrates a block diagram of a polarization matchingprocessor. As indicated in FIG. 6, receive signals Rx₁ and RX₂ are fedto an amplitude/phase estimation unit 602. The received signals are peruser and contain parameters that characterize a channel. In a GSMsystem, the channel parameters reside in equalizer tap weights. In aCDMA system, the channel parameters reside in a RAKE receiver fingers'information.

[0081] The estimates from 602 are then smoothed (i.e., longer timeconstants for slowly time-varying situations) at 606 using theinformation about channel dynamics that are estimated at 604. Thechannel dynamic estimates are based upon reverse link data. Theseestimates affect the time constants of the filtering performed on theRx₁ and Rx₂ receivers' snapshots. Channel dynamic estimates at 604 arefed to long term amplitude estimation filters 610.

[0082] The smoothed amplitude and phase information from 606 are fed toa quadrant detector 608 so that the sign of the relative phase can beestimated. This determines the quadrant (or pair of quadrants) of thetransmitted polarization. The smoothed amplitude information is also fedto a filter 610 that estimates the long term amplitude of thecorresponding reverse link. A time-varying magnitude (ratio between thetwo orthogonal polarizations) is based, in part, on the user-channeldynamics and the smoothed amplitude data. The time-varying magnitude isderived at 610 and is sent to a normalized weight generator 612 where apair of Transmit control weights α and β plus the sign from the quadrantdetector 608 are generated; thereby producing the complex weightW_(n)=α_(n)±jβ_(n) having a constraint: |W_(n)|=1, where the sign isdetermined from the quadrant detector 608.

[0083] In general, receivers employ a maximal-ratio-combining (MRC)diversity scheme, and an algorithm of a receiver MODEMdiversity-combiner to optimally combine the two received samples. Assuch, processor need not receive snapshots Rx₁, Rx₂ but should receivethe MRC (or other diversity combining scheme) parameters, namely theinstantaneous, which means per symbol, or per slot diversity combineroutputs, amplitude ratios and relative phase between the two receiverchannels. Thus, the “Amplitude/Phase Estimation (short-term)” block 602outputs the short-term amplitude and phase.

[0084] If the outputs of the amplitude/phase estimation block 602 arenot received from the receiver MODEM diversity-combiner, then theoutputs are determined as follows. First, the vectors Rx₁, Rx₂ areprocessed such as to consider only the un-faded portions of the vectors.As such, the data of the faded time-portions of the vectors (i.e.,≧F_([dB])below moving-average mean level, 3<F<7 a parameter) iseliminated. Then, the relative phase and amplitude of the processedvectors Rx′₁, Rx′₂ are estimated using the short-term (moving-average)complex normalized-covariance between the vectors Rx₁, Rx₂.

[0085] When the optimal polarization coincides with the crossedpolarized receive antennas, the phase detector output produces errors.Thus, the inputs to the “Quadrant Detector” (i.e., the two amplitudes,or rather their ratio) are discriminated against in boundary situationsto produce the correct phase.

[0086]FIG. 7 is a flow chart illustrating the method to controlpolarization. The method is applied to each user independently. Themethod begins at block 702 and control passes to block 704. At block 704a default vertical polarization is set (i.e., α_(n)=β_(n)={squareroot}2/2). The selection of a default vertical polarization is basedupon the fact that base stations employ a plurality of verticallypolarized antennae which in-turn complement the vertically polarizedantennae of most mounted mobile stations that are non-handheld. Afterthe default vertical polarization is initialized at block 704, controlpasses to block 706.

[0087] At block 706, polarization matching is performed as previouslydiscussed with regard to FIG. 6 to produce an estimated polarizationweight. Control then passes to block 708.

[0088] At block 708, it is determined whether the estimated polarizationweight results obtained in block 706 are consistent. If the results fromblock 706 are unstable or there are abrupt changes in the polarizationestimates, then control passes to block 704 to set the polarization tothe vertical polarization default state.

[0089] If the results are determined to be consistent at block 708, thencontrol passes to block 710. At block 710 the polarization of thetransmit antennas is rotated to match the estimated polarizationdetermined at block 706. Control passes to block 712 in which the methodto control polarization ends.

[0090] The information of each user may be compiled as an aggregate ofthe controls of all users, which may serve to evaluate fault performanceand management of the BTS. For instance, if a large proportion of theweights of the BTS vary abruptly for most or all of the users, it may beinferred that a moving object, such as a vehicle or a person, is inclose proximity to the BTS antennas.

[0091] Based upon the above embodiments, the following hardwareconstraints may be considered:

[0092] The antennas may contribute the following errors: (1)Non-coinciding phase centers for the two polarizations, in Rx and in Txwill induce an error that is dependent on the angle of arrival (AOA).Thus, care must be applied in the design of the BTS antennas to thephase centers for the two polarizations (Rx or Tx) so that they closelycoincide. (2) The relative gains of the two polarizations, in Rx and inTx, should track each other. Thus, the dB difference in gains betweenthe two Rx antennas, and between the two Tx antennas over the sectorangular span should be near 0 dB.

[0093] The relative gain response [dB] and relative phase response [°]between the two Rx paths, between the two Tx paths, and between the Rxpair and the Tx pair, is important for the successful operation of thepolarization matching processor. These parameters should to be knownover frequency and temperature.

[0094] The RF chains in Rx should track in amplitude and in phase, or becalibrated over temperature and frequency. A stored calibration tablemay then be used to correct for any phase and/or amplitude discrepanciesbetween the two Rx chains, and similarly for the two Tx chains.

[0095] A mobile unit which is located at some distance from the BTS witha dual polarization antenna, which reports its measurements of the pilotsignal (in a CDMA system) may be used to calibrate and validate theoperation of the system.

[0096] Upon switching the two cross-polarized antennas at the stationarymobile, the receptions of the pilot as well as the transmissions to theBTS will vary at the switching rate. The transmissions may be processedat the BTS to produce the polarization angle mismatch between thereverse and the forward link, and correct for it with a suitable unitarytransformation at the polarization matching processor.

[0097] In polarization matching, the polarization orientation intransmitting (per user) is adaptively determined based on the averageorientation estimated from the reverse link. FIG. 19 gives a high-levelblock diagram of polarization matching where the reverse link signal(per active user) received by two Rx cross-polarized antennas isprocessed to generate a pair of scalar weights (including sign). Theweights are then used to determine the relative gain of each of thetransmit forward link signals to be transmitted via the cross-polarizedTx antennas.

[0098] To implement polarization matching, there are many systemconsiderations such as power constraints. Since transmission lossescaused by polarization mismatch further affect other performance relatedmeasures (e.g., power usage, self-interference, orsignal-to-noise-plus-interference), implementations under differentconsiderations yield different overall performance. What is needed iseffective implementations for polarization matching, preferably in bothforward and reverse transmissions, that not only mitigates thetransmission losses but also performs to a maximum efficiency withrespect to various measures.

[0099] In system 1900, when the incoming signals Rx₁ and Rx₂ from areverse link are received from antenna pair 1915, a set of measurementsare adaptively estimated at 1905. Such measurements are related to bothfast and slow fading. Particularly, the average orientation of thecorresponding MS is estimated at 1950. Based on the measurements fromthe receive signals, a set of control parameters may be generated at1905. The control parameters include the parameters that are used toadjust the polarization orientation. The polarization control parametersmay be fed to an Rx polarization adjustment mechanism 1910 so that thepolarization orientation for the receive link can be matched with theaverage orientation. At the same time, the polarization controlparameters for the forward link can be sent to a Tx polarizationadjustment mechanism 1940 to re-orient the polarization of the transmitlink so that signals can be sent out over a pair of polarization-diverseantenna 1930 in an optimized polarization orientation.

[0100] In implementing the embodiments for polarization matching asdescribed above, there are many system considerations that mayultimately affect the performance. Examples of such considerationsinclude polarization diversity, power constraints, and choices ofbaseband/RF/IF implementations. Alternatives of polarization diversitycan be, for example, space-separated diversity, circular-polarization,or cross-polarization. For power constraints, one major issue related tosystems is the underlying Tx power constraints, in conjunction withpolarization matching and the BS power control methodology.

[0101] The present invention presents various implementation embodimentsfor both receive polarization adjustment mechanism and transmitpolarization adjustment mechanism under different system considerations.In particular, RF/IF and baseband system implementations are presented.Alternative implementations that satisfy different power constraints arepresented. The implementations of polarization matching in both forwardand reverse link are also presented. Finally, implementations ofpolarization that comply with different standards are illustrated.

[0102] Considering a forward link, given a desired transmit polarizationorientation (derived from the receive signals at the base-station fromthe same user), the transmit signal is to be sent at a slant angle offthe vertical reference orientation in the range of (−90°, 90°). To doso, the implementation of polarization matching requires twoorthogonally polarized antennas that simultaneously transmit tworeplicas of the transmit waveform, with a predetermined electrical phaseshift and relative gains between the two replicas. Usually the twoantennas are realized as cross-polarized ±45° linearly polarizedantennas.

[0103]FIG. 20 depicts polarization matching system 2000, constructed andoperative in accordance with the present invention. System 2000implements a baseband scalar weighting function to achieve polarizationmatching. In particular, system 2000 comprises baseband processor 2010,which produces two replicas of the transmit signal (in complex I-Qvectors). An appropriate complex weight is applied to one arm (relativeto the other). This weight (less calibration) represents an amplitudeweight plus sign. The two replicas are subsequently applied to a pair ofdual coherent (sharing a common local oscillator—(LO)) modulators and upconverters 2020 a and 2020 b. The modulated pair of replicas are thensupplied to a pair of linear power amplifiers 2030 a and 2030 b. Finallythe amplified signals are fed to the cross-polarized antennas 2040 a and2040 b in order to effect signal transmission.

[0104]FIG. 21 illustrates polarization matching system 2100, constructedand operative in accordance with another embodiment of the presentinvention. System 2100 implements a scalar weighting function in theRF/IF portion to achieve polarization matching. In system 2100, basebandprocessor 2110 supplies the transmit signal to modulator andup-converter 2110 because only one modulator and converter 2110 isrequired, which is then multiplied by the number of users. I and Qvectors of the up-converted signal are fed to RF/IF portion 2120 a,bwith an appropriate complex weight applied to one arm. The output of2120 a,b are then combined with other channels and fed to poweramplifiers 2130 a,b for transmission.

[0105] In the two implementation schemes noted above (as well as otherimplementation schemes discussed below), it is assumed that the totaltransmit power required is preserved or possibly reduced in somescenarios. Therefore, each of the two power amplifiers in theseimplementations should be specified to 50% of the power required by theoriginal single power amplifier used in a configuration with a singlevertically polarized transmit antenna.

[0106]FIG. 22 illustrates polarization matching system 2200, constructedand operative in accordance with another embodiment of the presentinvention. System 2200 incorporates both a baseband and an RF/IFpolarization matching implementation. In system 2200, a maximum powerrequirement per amplifier is guaranteed, both per user and per allusers.

[0107] As indicated in FIG. 22, two equal-power versions of a transmitsignal are generated at splitter 2210. Phase shifter 2220 shifts thephase of one of the two equal-power signals to create a phase differencebetween the two. The phase shift is performed at ‘low power’ (i.e.,baseband—preferably, or RF, or even IF, if one prefers to use a dualcoherent up-converter). These two equal level signals are then appliedto the power amplifiers 2230 a and 2230 b. Thus, a maximum utilizationof the two amplifiers is guaranteed. The 0/π hybrid 2240 produces, asits outputs, both a sum and a difference of its inputs (the twoamplified equal power signals). Even though these two outputs are notnecessarily of equal amplitudes, they are always orthogonal to eachother. Finally, phase shifter 2250 imposes a 90° shift to align the twosignals back before they are applied to the cross-polarized antennas2260 a and 2260 b.

[0108] The splitter 2210 and the phase shift 2220 must be performed on aper-user basis. They may be performed at baseband or IF or Low-Power RF.All of the users' two channel (equal power) outputs are summed in eachcorresponding arm before entering the power amplifiers as well as thedual-polarized antennas.

[0109] Since the transmissions from all users are equally split betweenthe two arms and the two equal-power signals are uncorrelated, the sumsacross all users in the two arms are also equal. This realizationmethod, therefore, guarantees a full loading of the two power amplifiersfor any polarization Matching setting with respect to any user.

[0110] The phasor diagram in FIG. 23 illustrates the fundamentals of theoperations described above. Assume that the input vector is split intotwo equal vectors, one of which is rotated by a phase shifter. Atbaseband, this rotation operation may be performed via a 2×2 matrixoperation (sometimes referred as CORDAC). It is basically a scalaramplitude plus sign (calibration is not included). The resulting (equalsize) phasors are denoted by A and B in FIG. 23. The vector representingthe sum of A and B is denoted by A+B and the vector representing thedifference of A and B is denoted by A−B. The sum vector A+B bisects theangle formed by vectors A and B. The different vector A−B bisects theangle formed by A and −B. Since the two angles formed by AB and A(−B)(each belonging to a different rhombus) complement each other to 180°,the angle between the sum vector A+B and the difference vector A−B is90°. A continuous wave (CW) rotation of the sum vector (or thedifference vector) by 90° co-aligns the two vectors, producing an linearpolarization with the orientation determined by the ratio between themagnitudes of the sum and the difference vectors. This ratio isdetermined by the amount of phase shift between A and B.

[0111] It is possible to use a different phase shift, instead of a shiftwithin a fixed range (between 0° and 180°), to get −A to produce bothvectors A+B (as before) and −A+B. In this way, vector −C, instead of C,can be obtained after a CW rotation. Therefore, with a (0°-360°)variable phase shifter, together with a hybrid and a fixed 90° shifter,it is possible to achieve any linear polarization orientation. Using thecosines equation, it can be demonstrated that for polarization angle ψ,which is a function of the electrical phase shift φ, the followingholds:$\psi = {{tg}^{- 1}\sqrt{\frac{A^{2} + B^{2} - {2\quad {AB}\quad \cos \quad \varphi}}{A^{2} + B^{2} + {2\quad {AB}\quad \cos \quad \varphi}}}}$

[0112] When A=B, the following equality can be derived:

[0113] This equality means that the polarization slant angle ψ is onehalf of the

ψ=φ/2

[0114] electrical shift φ. As for the resultant polarization slantedvector, its size will be {square root}2*A regardless of the value ofφ(or ψ). Therefore, the resultant power is the sum of the two outputs ofthe two power amplifiers (per user).

[0115] It is commonly known that when a vertically polarizedtransmission from a BS results in a polarization mismatch, highertransmitting power is required to compensate the polarizationtransmission losses. The increased transmitting power subsequentlycauses increased self-generated interference. In the illustratedembodiments of systems 2000 and 2200 (FIG. 20 and FIG. 22,respectively), such polarization mismatch losses are recovered bydriving the power amplifiers at non-equal vs. equal power levels, peruser.

[0116] The implementation of system 2200, where two equal powercomponents (per active user) drive the two power amplifiers, guaranteesa uniform and efficient utilization of the amplifiers. Such a uniformutilization of the amplifiers is also guaranteed for a collection ofactive users. This desirable characteristic is due to the fact that theunequal components that constitute the slanted polarization vector aregenerated after the power amplification (by the 0/π hybrid at 2240).Even though the power levels to the amplifiers may vary from one user toanother due to the different underlying power control requirements atthe BS, the inputs to the pair of power amplifiers 2230 a and 2230 b atany given moment are always at an equal power level. As a comparison,the polarization matching implementations used in systems 2000 and 2100where each amplifier usually gets (per user) non-equal power levels, theunequal components that constitute the slanted polarization vector aregenerated before the power amplification stage.

[0117] Loading of power amplifiers is one of the main considerations inCellular/PCS transmission systems and is directly related to theperformance of the linear power amplifiers. There are differentalternative approaches to realize power control. One approach is tocontrol the total transmitted power by an instantaneous peak-powerlimitation. A different approach may be to specify a constraint on themaximum total mean power out of the linear power amplifier. Yet anotherpossible approach is to constrain the power transmitted per active userunder a fixed total mean power constraint employed at a BS.

[0118] The implementation of polarization matching per active user mayemploy one of the two power constraint rules described below. One ruleis to have the two weights sum up to 1 (as indicated in FIG. 6). In thismanner, when the required polarization matches one of the two Txcross-polarized antennas, the weight is 1 for the transmitting antennaarm that matches the required polarization orientation and the weightfor the other is obviously 0. For any other polarization orientation,the two weights (per active user) are attenuated such that the magnitudeof the resultant polarization is still 1. This amounts to a fixed powerconstraint per active user.

[0119] A different rule may also be applied. Instead of summing to 1,the sum of the two scalar weights may be greater than 1. The two scalarweights are managed such that at least one of them is 1, and the otheris less than or equal to 1. When the required polarization is alignedwith one of the cross-polarized Tx antennas, the weights will be 1 and 0as in the first rule. When the required polarization orientation is±45°, both weights will be set to 1. For any other polarizationorientation, one weight (the larger of the two) will be set to 1, whilethe other will be less than 1. This type of power control does notpreserve the total power but guarantees a pick power limit for each arm(per active user).

[0120] It is assumed here that, due to emission requirements, thereexists a constraint that limits the maximum mean power at the output ofthe power amplifier. Such a constraint is usually also associated with arelated peak power constraint. For example, a combined constraint can bethat the ratio between the peak and the mean power obeys some limit(e.g., peak-to-mean=10 dB).

[0121] The amplifiers' loading situation related to the three differentimplementations of systems 2000, 2100, 2200 for polarization matchingare analyzed in FIGS. 24(a)-24(c), respectively. These three cases are(1) vertically polarized transmission illustrated in FIG. 24(a), (2)cross-polarized transmission with the weighting applied prior to thepower amplification (FIG. 24(b) or the implementations illustrated inFIGS. 20 and 21), and (3) cross-polarized transmission with theweighting applied after the power amplification (FIG. 24(c) or theimplementation illustrated in FIG. 22). The required power is denoted byP and the required polarization orientation is denoted by θ. Assuming acertain probability distribution of θ and a certain distribution of therequired power P, the loading of the power amplifiers for the threecases are presented.

[0122] The reference loading at the output of the amplifier for thefirst case is P, whereas for the second and the third cases (“half size”amplifiers), the reference is ½P.

[0123] In the first case (FIG. 24(a)), there is a significant overloaddue to the (cos θ)⁻¹ factor. Based on various statistical assumptionsabout the probability distribution of θ, overload of 5 dB per user (overP) may easily occur. In the second case (FIG. 24(b)), there is apossible overload due to the factor 2 sin²(45−θ) (or similarly with 2cos²(45−θ))) of up to 3 dB, per user, over P/2.

[0124] With k users, the probability of exceeding the specified maximummean power may decrease, depending on the statistics of P and θ per userand the number of users k. Usually, to avoid the overload of the poweramplifiers, a peak (total) power limit is enforced at the base-station.This means that some users will be deprived the power they need tosustain quality communications, even though it may be only a few percentout of the total population of served users.

[0125] In the third case (FIG. 24(c)), each power amplifier guaranteesto provide exactly P/2 and thus the power amplifiers are fully andevenly loaded. This is the most efficient case of all three. Such powercontrol is maintained by phase only, and is performed (practically) peruser at the baseband.

[0126] FIGS. 25(a-c) shows the simulation results with respect to thediversity gain in above described three cases. In the simulations, thepower requirement P varies uniformly between 0 dB to −10 dB and thepolarization angle θ around the vertical polarization orientation (0°)follows a normal distribution N (0°, 17°). Such simulation conditionsmay represent a single user or a collection of users. The plots in FIGS.25(a), (b), and (c) correspond to the three cases, individually. In eachplot, the horizontal axis is the relative required power in dB, with 0dB to be the maximum allowable power out of the amplifier. Severalimportant observations can be made from these plots.

[0127] One observation from those plots is that in configurations (a)and (b) there is an overload probability greater than 0 (e.g. ˜6% in(a), and ˜10% in (b)), whereas in (c) there is no overload. Anotherobservation is that in (a) the power spread is from an overload value tonot less than −10 dB. In case (b) it is between an overload value andsome −22 dB, whereas in case (c) it is strictly between 0 dB and −10 dB.

[0128] Yet another observation is that the distribution in FIG. 25(a)(corresponding to the first case) has a broad (but not uniform) shape;in FIG. 25(b) (corresponding to the second case), it is more peaked(although within 0 dB to −10 dB very similar to FIG. 25(a)); in FIG.25(c) (corresponding to the third case), it is uniform, similarlydistributed as the power requests.

[0129] The implementations presented in systems 2000, 2100, 2200 realizepolarization matching in the forward link. As for a reverse link, threeimplementations, systems 2700, 2800, 2900, illustrated in FIGS. 27, 28,and 29 allow equal power reception for any polarization orientation toachieve maximum diversity gain.

[0130] It is known that diversity reception may be implemented usingeither space-separated antennas (both vertically polarized) orcross-polarized (±45°) antennas. Applying maximal-ratio-combining (MRC),the resulting (scalar mean power) SNR equals the sum of the SNR's ateach branch. A highest diversity gain can be achieved when the twoantennas receive the same power. With cross-polarized antennas, it ispossible that the input from one branch have a higher SNR while theother has a much lower SNR. For statistically fading receptions, themost important factors that determine the diversity gain are thecorrelation coefficient ρ and mean power difference Δ (in dB) betweenthe two antennas. That is, the diversity gain varies with the mean powerdifference Δ. Since two cross-polarized antennas are located at ±45°, Δmay vary between 0 dB to 3 dB. The plot in FIG. 26 shows the effect ofboth Δ and ρ on diversity gain (FIG. 26 is based on the result reportedin “An Experimental Evaluation of the Performance of Two-Branch Spaceand Polarization Diversity Schemes at 1800 MHz”, authored by A. M. D.Turkmani, A. A. Arowojolu, P. A. Jefford, and C. J. Kellett, IEEETransactions on Vehicular Technology, Vol. 44, No. 2, May 1995, pp.318-326). In FIG. 26, the X axis represents Δ, ranging from 0 dB to 10dB, and the Y axis represents the diversity gain. Different curves inFIG. 26 correspond to different values of ρ, ranging from 0(corresponding to the upper most curve) to 1.0 (corresponding to thelower most curve) with step size 0.1. It is clear in the plot that thebest diversity gain can be achieved when Δ=0 dB. That is when the twoantennas receive the same power. This is also true even when the poweron each antenna is less by 3 dB than the full power on just one of thetwo antennas (with nothing on the orthogonal antenna).

[0131]FIG. 27 illustrates another embodiment, a polarization matchingsystem 2700. The equal (half) power may be possibly guaranteed in thetwo branches with respect to any polarization orientation. This rendersthe MRC weight into a phase shifter, with improved relative phaseestimation due to the guaranteed (3 dB less than the maximum) high SNRon both branches.

[0132] In system 2700, the outputs from the two cross-polarized antennasare fed into a 90° hybrid 2720, which produces two orthogonal outputsfrom circularly polarized antennas. After splitting into multiple MODEMs(one per channel/user), the MRC algorithm is operated on the twobranches to yield an enhanced resultant output.

[0133] System 2800 presents another embodiment of the invention thatguarantees to produce two equal power vectors from the two (generally)unequal vectors intercepted by the cross-polarized antennas. Theimplementation of system 2800 is similar to what is implemented forrealizing polarization matching in a forward link.

[0134] Another embodiment is illustrated by system 2900. In somerespects, system 2900 is similar to system 2800 except somemodifications on the locations of the low-noise power amplifiers. Oneimportant characteristic of system 2900 is that the entire processingnetwork may be realized at baseband after coherent dual-channeldown-conversion is performed using any diversity scheme.

[0135] The polarization diversity gain is a function of parameters Δ andρ, as described before. In reality, since the power of the two antennaschanges with to the orientation angle of the polarized wave, parameter Δvaries. In addition, the correlation ρ between the two receptions alsodepends on the orientation angle. Consequently, the polarizationdiversity gain fluctuates with the orientation angle as well. Thisbehavior is depicted in FIG. 30 where the X axis represents the valuesof the orientation angle and the Y axis is the diversity gain. Two ofthe three curves in this figure (the ones marked as “Without Network”and “With Transform Network”) illustrate the behavior. Therefore, withsystems 2800 and 2900, even though Δ=0 dB is achieved, the systems maybehave non-optimally compared with what is described in FIG. 26.

[0136] It is observed that for some orientation angles, system 2700yields better polarization diversity gain than systems 1100 and 1200.But for other orientation angles, systems 2800 and 2900 yield betterpolarization diversity gain than system 2700. It is, therefore,desirable to form another embodiment in which two different processors(one derived from system 2700 and the other derived from 2800 or 2900)are combined at the baseband. Better linear pre-processing can beapplied before the two signals are fed to the maximal-ratio-combiner(MRC). By doing so, the combined system always yields the best possiblediversity gain. The curve marked as “Maximum of Gains” in FIG. 30demonstrates such optimized behavior.

[0137] The implementation schemes presented so far may be applied tomany full duplex wireless communication systems, with differentmultiple-access regimes, including the well known TDMA and CDMAstandards. Below, the feasibility of applying the present invention toseveral representative standards is discussed.

[0138] According to TDMA: GSM standard, the waveforms occupyinstantaneously 200 KHz in each link. The signals suffer from fading inaddition to polarization mismatch losses. In a system complying withthis standard, the polarization orientation does not vary significantlywithin a single frame. Therefore, it is possible to apply polarizationmatching to improve the reverse link performance at a gain of severalnet dBs. The BS employs one transceiver per channel (frequency),carrying 8 time division multiplexed calls, with one power amplifier perchannel. It is necessary to split the low power RF into two equal powerports, to phase shift one of them, and then to power amplify using twopower amplifiers each of which is rated at one half of the power of theoriginal amplifier. After the amplification, high power 180°-hybrid plus90° delay mechanism at the outputs of the power amplifiers can beapplied to get the polarization matched transmission per active slot(per user). It is to be noted that the phase shifter at the low power REmust be set to an optimal value for each user. Therefore, it changesfrom slot to slot.

[0139] The implementation of system 2200 with low RF input may be usedin a system that complies with TDMA: GSM standard. For a multi-carrierGSM system, each carrier should be processed separately as describedabove and the two outputs into the cross-polarized antennas per eachcarrier should be combined to yield a single multi-carrier output intoeach of the two cross-polarized antennas. In cases where the output froma high power 180°-hybrid plus 90° delay may be realized with enoughflatness and power handling capability for the full bandwidth of themulti-carriers, it is possible to reduce the number of these networks.The reduction may be achieved by summing the outputs of the poweramplifiers with respect to polarization orientations (separate theoutputs destined to different polarization orientations), respectively,and then feeding the combined outputs into a single high power180°-hybrid plus 90° delay unit.

[0140] According to standard TDMA: IS-54/136, waveforms occupyinstantaneously 30 KHz in each link and signals suffer from severefading in addition to polarization mismatch losses. Since the meanpolarization orientation should not vary abruptly over 20 msec (which isthe length of one time slot), it is possible to apply polarizationmatching to improve the reverse link performance at the level of severalnet dBs. The hardware realization depends on existing BS architectures.Commonly employed architectures use one transceiver per channel(frequency which carries up to 3 TDMA calls) with one Tx port and two Rxports at low-power RF. The interface between the ports and the antennasis usually through combining networks (lossy) with, often, multi-carrierfeedforward amplifiers.

[0141] To deploy polarization matching, a vertically polarized antennamay be replaced with a dual-polarized antenna. The power amplifier maybe modified in such a way that the equal total power is ensured for thetwo antennas (one half per antenna), with two inputs and two outputs perchassis. Thus, only the amplifiers' input splitter and output combinerneed to be changed. The DC supply rails are applied to each half of anumber of plug-in power amplifier modules, all of which belong to thechassis. The power amplifier modules, including the number of suchmodules, need not be modified at all. The additional block required inthe low power RF (in the Tx path) section is a non-uniform and variablepower splitting network, one per frequency. Each network consists of one1:2 power-splitter and one phase shifter per channel, two (instead ofone) combining networks into the two power amplifier inputs, and onehigh power 180°-hybrid plus 90° delay at the outputs of the poweramplifiers. This corresponds to the low-power RF implementation ofsystem 2200, described in FIG. 22.

[0142] Current 2G CDMA systems (complying with CDMA:IS-95) employ asingle modulator and up-converter that transforms the summed complexbaseband signals into a single carrier transmission. To applypolarization matching, it is required to modify the baseband ASIC and toperform per user (or per active call) splitting (into 2 Tx complexbaseband and phase shift one complex arm with respect to the other). Inaddition, the ASIC supports dual summation of the complex basebandoutputs from multiple users with respect to each destined polarization(there are two orthogonal polarizations). Another modulator andup-converter unit is added to the sector equipment with common localoscillators (LO's) between the two parallel units. The two summedbaseband complex outputs are then fed into two coherent modulators andup-converters. The low power dual summed RF signals are entered into adual multi-carrier amplifier with two inputs and two outputs (could bethe same multi-carrier amplifier currently deployed with a redesignedchassis to assign half of its modules to one polarization and the otherhalf to the other polarization). The high amplifiers' power outputs arefed into a high power 180°-hybrid plus 90° delay and finally to adual-polarized antenna. This completes the transmission chain percarrier.

[0143] The CDMA: 3G standards (WCDMA, CDMA2000) employ some new andimportant features in the forward link that are most relevant to thepresent invention. They address transmit diversity using S-T coding withtwo transmit antennas. Conceptually, the two transmissions are from thesame information data but with related yet different coding. These twotransmissions are generated and power amplified prior to beingtransmitted via two separated antennas that are both verticallypolarized and removed horizontally from each other by 10 to 20wavelengths. This is a form of space-diversity in Tx. The enhancement isachieved because of the coding employed. The transmissions from all theusers (per approximately 4.75 MHz carrier, or 3×1.25 MHz) are summedwith respect to each transmit antenna and then amplified using a linearpower amplifier of the corresponding antenna.

[0144] A 3G system may benefit from polarization matching by employingit in each of the two transmit chains separately. The improvement ofseveral net dBs in E/(I+N) may be obtained in addition to any other typeof gain. To do so, the S-T coded transmission in a 3G system mayco-exist with the polarization matching scheme.

[0145] Specifically, the 3G-CDMA ASIC at the BS may employ (1)polarization diversity reception (MRC) for every dual polarized antennaand (2) space diversity for every pair of space-separateddual-polarization antennas. Therefore, a total of 4 Rx diversities (2for polarization diversity and 2 for space-separated diversity) may beemployed with two imtermediate combining weights (1 for eachcross-polarized antenna). From the MRC of every cross-polarized antennaat the receiver end, the polarization matching Tx orientation may beestimated. During transmit, the ASIC should provide the splitting plusrelative phase-shift capabilities for each user (or each active call)per output (there are 2 S-T coded outputs per user), which is similar towhat is described for the IS-95 CDMA case. The 4 summed outputs (2 forevery space-diversity antenna) are modulated and up-converted using 4coherent chains (i.e. all using one common set of LO's), as opposed to 2in the basic 3G transmit architecture.

[0146]FIG. 31 depicts system 3100, constructed and operative inaccordance with another embodiment of the present invention. System 3100provides polarization matching scheme for a 3G-CDMA implementation. Asindicated in FIG. 31, system 3100 comprises four linear poweramplifiers, 3130 a,b and 3170 a,b, to amplify the low-power RF combinedsignals. Each pair of power amplifiers 3130 a,b, and 3170 a,b is for onepair of space diversity antennas (with polarization matching), 3150 a,b,and 3190 a,b and will have a combined mean power rating which equalsthat of the (single) power amplifier per vertically polarized antenna.Finally, two pairs of high power 180°-hybrids 3140, 3180 plus 90° delayswill interface the two pairs of the outputs, from power amplifier 3130a,b and 3170 a,b, into the corresponding pairs of cross-polarizedantennas. The power sum of the four amplifiers 3130 a,b and 3170 a,b iseither the same as what is required in situations where just twoamplifiers are employed in the basic S-T coding scheme or less, due toincreased power efficiency resulted from the polarization matchingimplementation.

[0147] Due to various limitations, it may not be possible for some 3Gsector implementations to support the space diversity configurationrequired for S-T coded transmission. In such situations, one option maybe to deploy cross-polarized antennas in the limited space allocated. Assuch, the reverse link may be operated by the MODEM with polarizationdiversity reception, similar to that in IS-95 systems. In the forwardlink, however, the scheme of S-T coding with transmit diversity requiresthat the two paths (each with independent fading), from the antennas(BS) to the user (MS), will be of comparable mean power. Applying thepolarization matching to each user will yield the estimate for the meanpolarization orientation of each user. Using such estimated meanpolarization orientation, the two S-T coded signals (per user) may betransmitted in the polarization orientations that are +45° and −45° offthe estimated mean polarization orientation.

[0148] Consistent with this notion, FIG. 32 depicts system 3200 thatprovides such as implementation. In system 3200, the sector transmitstwo signals, each attenuated by 3 dB with respect to the fully matcheduser polarization, but with equal mean power and independent fading,thus enabling the S-T coding to function efficiently.

[0149] Another possible alternative is to transmit the two versions ofthe S-T coded signals in two orthogonal circular polarizations (CW andCCW). This may be realized by using a phase shifter of less than π/2between the 180°-hybrid and antenna in the scheme of system 1400. Thesame 3 dB loss will occur. Using the RAKE receiver and the S-T decoder,the user at the receive end can benefit from the diversity.

[0150] The following embodiment describes the extension of polarizationmatching to a multiple antenna elements case. A multiple arrangement ofantenna elements is called an antenna array. When the elements arearranged in a vertical array, it is referred to as a column array. Whenthe elements are arranged in a two-dimensional array, they are usuallyconfigured as a multi-column array. In the following discussion, thepolarization matching operation in arrays is described.

[0151] The array antenna radiation pattern in receive is archived viaweighting the signal at each array element (relative amplitude andphase) and combining of the weighted array element signals into thearray beam output.

[0152] In transmit, the signal at the array beam port is divided intoseveral equal outputs, each weighted (relative amplitude and phase) andfed into the transmitting elements.

[0153] Generally, applying a set of weights to an array will result inan antenna pattern, or beam, and is called beamforming. A special caseof beamforming is beam steering. In the case of beam steering, thearray, which generally is equispaced in the vertical dimension, andequispaced in the horizontal dimension (each possibly different from theother), forms a narrow well-defined beam pointing in a certaindirection.

[0154] The relative phases along the elements in each dimension(horizontal and vertical) form an arithmetic series (vary linearly alongeach dimension) when the elements are uniformly spaced in each axis(vertical, horizontal). Steering the beam in each dimension requiresjust a single parameter per dimension in this case, the “steeringangle”.

[0155] When transmitting and receiving is performed with an antennaarray, the polarization matching methods described above are extendedinto combined polarization and beam matching.

[0156] An algorithim can be applied to an antenna array in which thesignal is received via the multiple-elements array, using an array ofdual slanted polarization antenna elements. Upon reception, the steeringangles (in azimuth and/or elevation) that result in maximumsignal-to-noise power ratio are found, while the outputs of the twoslant polarization arrays (e.g. +45° and −45° polarizations) arecombined to achieve an output which is maximum in signal-to-noise ratio.

[0157] One such algorithm is maximal-ratio-combining (MRC), which wasdiscussed in detail above. The relative weights of the two slantpolarization arrays' outputs provide the information that determines themean polarization of the received signal across the array.

[0158] Based on the received beam best polarization and steering angleestimation, a transmission to the same user can be performed by steeringthe transmit slant arrays to the same direction and orienting thetransmit polarization according to the mean receive polarization asexplained above.

[0159] Practically (and in general) the dual polarized antenna arraycharacteristics will be such that it is not guaranteed that the phasecenters of dual polarized antenna arrays will be co-located. For thisreason, the polarization matching algorithm that was described for asingle element with co-phased cross-polarized antennas may be extendedinto a two-dimensional array and polarization-matching algorithm.

[0160] In this extended algorithm the relative amplitudes and phases ofthe array elements at each polarization (of the two orthogonalpolarizations) are modified in such a way that two objectives arefulfilled:

[0161] 1) The effective polarization per cross-polarized elementresulting from weighting the two orthogonally polarized antennas(realized in an element) upon Tx should match that same user receivedmean polarization.

[0162] 2) The relative phases and amplitudes of the array pairs ofpolarization matched elements (after weighting and 2:1 combining to formone “polarization agile” element) should be such that the overall arrayresulting pattern will point at that user with the maximum gainpossible.

[0163] In this embodiment, polarization matching in transmit is based onthe mean polarization in receive, and both polarization and array beammatching are combined for this purpose. The extended algorithm presentedabove can be applied to an algorithm that performsmaximal-ratio-combining (MRC), whereby all complex signals into or outof the array elements undergo complex weighting to achieve the bestsignal-to-noise power ratio. The MRC algorithm was described in greaterdetail above.

[0164] The method presented above in the case of the generaldual-polarized array can also be described as forming in transmit, bothin the polarization and array pattern spaces, a retro-directive beam,based on the mean receive optimum polarization, and employing arraybeamforming matched to the specific user transmission.

[0165] In addition to the general case presented above, it is importantto note several special cases, where the array has a well-controlledphase-center matching for the two orthogonal polarizations.

[0166] For a single column array the beam steering is performed only inelevation, this is called tilting the beam. Thus, we have in this casejoint polarization matching and beam tilting. For a multi-column array,both tilting and azimuthal steering are applied for each of the twoslant-polarization arrays, accompanied by polarization matching.

[0167] One implementation of receive array beam steering andpolarization matching is via sequential/staged optimization: first, beamsteering is performed on each of the two slant arrays, followed bycombining for matched polarization reception.

[0168] Another implementation is to optimally weight 2N-element array,where N elements are in one polarization, and the other N elements arein the other (orthogonal) polarization.

[0169] If the 2N-complex vector of signal envelopes received at thearray elements is denoted by S, then the optimum weight will be:$W = \frac{S^{*}}{S}$

[0170] Where the asterisk denotes a complex conjugate, and the vector Wis normalized. It is assumed in the equation above that all elementshave additive noises, which are normal zero mean and i.i.d. (i.e.,independent identically distributed).

[0171] The orthogonality of the two polarizations leads to theconclusion that the entries of the vector S can be arranged in such away that it can be expressed as: $S = \begin{bmatrix}S_{1} \\S_{2}\end{bmatrix}$

[0172] where S₁ and S₂ and n-dimensional vectors each, corresponding tothe two polarization-arrays. The weight vector can be expressed ascorresponding weights per each array: $W = {\frac{\begin{bmatrix}S_{1}^{*} \\S_{2}^{*}\end{bmatrix}}{S} = {\frac{\begin{bmatrix}S_{1}^{*} \\S_{2}^{*}\end{bmatrix}}{\sqrt{{{S_{1}}^{2} + {S_{2}}^{2}}}} = \begin{bmatrix}W_{1} \\W_{2}\end{bmatrix}}}$

[0173] It can be shown that this set of weights performs both beamsteering and polarization matching as described above.

[0174] The viability of polarization matching of the above embodimentswere confirmed during field trials which are described below. Inparticular, the field trials confirmed the correlation between thepolarization vectors of the forward and reverse links. The trialincluded the following:

[0175] Transmission of both links from the MS, and reception at the BTS.This simplifies the test equipment and procedures considerably. Theresults are valid, by recognizing that the channel is reciprocal for anygiven frequency.

[0176] The transmit antenna—a linear dipole, is rotated in the verticalplane, thus providing a periodically variable polarization for bothfrequencies.

[0177] A linearly polarized receive antenna at the BTS receives signalsthat fluctuate periodically in time.

[0178] The cross polarization discrimination (XPD) is measured fromthese fluctuations.

[0179] The correlation between the average polarization vectors (thepolarization ellipse) of the two frequencies is then measured.

[0180] The multipath fading is measured for a non-rotating transmitantenna, and the correlation between the fading in the two frequenciesevaluated.

[0181] The trials were conducted at PCS frequencies, LOS and NLOS, Ruraland Sub-Urban, and at varying speeds (0 to 70 Km/hr). As part of thetests, a mobile transmitter station and a stationary receive stationwere used. The tests confirm polarization matching between thefrequency-separated transmissions, both for LOS and for highly clutteredNLOS channels.

[0182] The test transmitted over a wireless mobile channel varyingpolarizations, at two distinct frequencies, in order to simulate theup-link to down-link FDM separation. Because of the reciprocal nature(with respect to the channel) of the long-term average polarization, thetransmit was performed from one site and the receive from another. Inparticular, the mobile site was chosen as the transmitting site, whilethe receiving site was stationary.

[0183] The two PCS continuous-wave (CW) sources, which represented theup-link and down-link frequencies, were transmitted through a linearlypolarized antenna. The rotation of the antenna in the vertical planecauses a varying transmit polarization. Thus, scanning all polarizations(vertical to horizontal) enables the evaluation of the channel XPD atboth frequencies upon reception by a linearly polarized antenna, as wellas to compare the relative reception of the two frequencies over time.Polarization matching exists if highly correlated average power isreceived at the two distinct frequencies.

[0184] The range of the test area was between 2-5 Km, and in rural andsub-urban environments, both LOS and NLOS.

[0185] In the trial system, the transmitter used two PCS CW synthesizedsources, at frequencies 1875 MHz and 1975 MHz. The combinedsynthesizer's outputs were power amplified into 20 dBm total mean powerin two sets of tests, and into 33 dBm total mean power in the third setof tests. The higher power was made available for the NLOS tests in asub-urban environment. The transmitter antenna was a dipole with betterthan 20 dB XPD. The whole transmitter assembly, including battery powersources were mounted on (and behind) a circular rotating plate thatserved as a (parallel) back plane for the transmitting dipole. Thedipole was mounted at the center of the plate, and transmitted to ahemisphere. The transmitter equipment was mounted on top of a roof of avehicle, and transmission was towards the rear hemisphere of thevehicle. The rotation caused the transmitted signals to attain alllinear polarizations in the plane of rotation, simultaneously, throughthe same antennas and into the same channel. In addition, the equipmentmounted on the vehicle was tested at a stationary position prior to arun of tests, to confirm the overall operation of the trial set-up. Thestationary receive equipment was placed at the antenna range controlroom and the linearly polarized receive antenna could be rotated both inazimuth and polarization.

[0186]FIG. 8 illustrates the test system layout for the trial tests. Thelinearly polarized antenna received the transmissions with a “fullrectified” shaped detector (mean-power) waveforms, reflecting thepolarization projection onto the receiving antenna according to:

ΔPower=20·log₁₀(abs(cos Θ)), [dBm]

[0187] At a rotation rate of 30 rpm there is one polarization “cycle”(from fully aligned to orthogonal) per second.

[0188] The receiving system comprised a linearly polarized patch antennafollowed by an LNA and split into two spectrum analyzers that served asselective receivers. The spectrum analyzers were operated in zero spanmode, with 10 KHz resolution bandwidth, 3 KHz video bandwidth, sweeprates variable from 100 milliseconds/screen to 20 seconds/screen. Thespectrum analyzers were operated under HPIB control that synchronizedthe sweep starts of the two spectrum analyzers, and the screen downloadsinto the PC (with 401 samples per screen per spectrum analyzer downloaded).

[0189] The technical specifications of the trial equipment were asfollows:

[0190] Antennas

[0191] 1 Tx Antenna: Dipole on a backscreen, Linear Polarization, XPD>20dB

[0192] 1 Rx Antenna: Huber-Zunner Patch, Linear Polarization, XPD>25 dB

[0193] Tx Polarization Rotation

[0194] Dipole rotated in vertical plane

[0195] Rate—One (1) rotation per 2 sec. (30 rpm)

[0196] Frequencies

[0197] 1× Carrier at 1875 MHz

[0198] 1× Carrier at 1975 MHz

[0199] RF Power

[0200] Tx Power—20 dBm, or 33 dBm total mean Tx power into antenna

[0201] DC Power

[0202] 1×12V/60 Ah Battery for assembly rotation

[0203] 1× or 2×12V/7 Ah Batteries for RF transmitter

[0204] Vehicle Mounting

[0205] Roof-top, wooden mounting floor

[0206] Plastic wind-shield (over a wooden enforcing structure)

[0207] Rotation on-off switch (from Battery) near the Driver

[0208] Receiving Equipment

[0209] LNA+2:1 power splitter

[0210] 2× Spectrum Analyzers (S/A operated under HPIB at Zero Span, withSynchronized Sweep Start, and Synchronized Sweep Download into PC)

[0211] 1× Oscilloscope, dual channel (to present S/A Aux. Videooutputs).

[0212] 1× PC

[0213] HPIB Cables, RF Cables, DC Cables

[0214] Communication

[0215] 2× Cellular Phones

[0216] This first set of the trial had the following specificobjectives:

[0217] 1. Generally confirm the Polarization Matching Phenomenon.

[0218] 2. Check and evaluate Correlation between Linear Polarizations(Projections upon reception on a Linear Test Antenna) of waveforms power(long-term average) at two distinct frequency bands over same WirelessMobile Channel.

[0219] 3. Test at PCS frequencies with realistic frequency separation,both LOS and NLOS, Rural and Sub-Urban, and under Varying Speeds (0 to70 Km/Hour).

[0220] 4. Confirm Sizeable XPD's of Received Signals in variousscenarios.

[0221] 5. Quantify the Power Value of Matched Averaged Polarization, asfunction of Scenario.

[0222] In addition, confirmation was made of well-known phenomena, suchas correlated (long-term) behavior of path-loss (including shadowing),with uncorrelated fast fading between the two signals, separated infrequency by 100 MHz.

[0223] The tested and/or controlled parameters relevant to the trialwere as follows:

[0224] XPD of Test Antennas (was measured);

[0225] XPD of wireless channel (LOS, NLOS, rural, sub-urban);

[0226] Polarization Matching between two distinct frequencies(correlation);

[0227] Fading of each signal, and their lack of correlation;

[0228] Mobile speed, environment; and

[0229] Reference fixed polarization measurements.

[0230] In the trial results, the two graphs represent the receivedsignals of two distinct frequencies, over time measured in a number ofsamples. The pre-tests were run after the installation of the equipmenton the vehicle, and setting up the trial system. The measurements weretaken while the vehicle was stationary at approximately 50 meters fromthe Rx antenna. FIG. 9 shows the effect of the rotating polarization,and the tracking of both signals. It also shows that an overall XPDlarger than 15 dB is exercised by the trial equipment.

[0231] The test drives were run at various speeds, with fixed (verticaland slanted) and rotating polarizations. The route was mostly LOS withsome NLOS sections. FIG. 10 shows the rotating polarization of thetransmit antenna, and the excellent tracking of the signals at twodifferent frequencies upon reception. Although occasional reflectionscause some fades, the channel does not suffer from heavy multipath. TheXPD preserved along this channel is at least 10-11 dB.

[0232] The suburban drive routes enabled testing with varying degrees ofreflections and multipath. In the LOS suburb, the route was up-hill withbuildings, which causes background clutter, as shown in FIG. 11. FIG. 11shows the rotating polarization of the transmit antenna, and theexcellent tracking of the signals at two different frequencies uponreception. Although occasional reflections cause some fades, the channeldoes not suffer from heavy multipath. The XPD preserved along thischannel is at least 10-11 dB.

[0233] In the NLOS suburb, the route was chosen such that the vehiclewas completely at NLOS with respect to the antenna range, and surroundedmostly by buildings. The received signals suffered a highly reflectiveenvironment, resulting in increased fading. FIG. 12 illustrates that inthis area the channel is generally NLOS with a narrow “window.”

[0234] Fading is exercised (independent on the two signals) with XPD of(at least) 10 dB, and a remarkable pattern of the rotating polarization.Again, the two signals track (long-term) each other in accordance withthe polarization matching.

[0235] The trial results clearly demonstrate that in rural, sub-urban,LOS and NLOS channel cases, an XPD of at least 10 db, following thecos²(0) law will occur. In other words, polarization mismatch willresult in a significant power loss. The tests also demonstrate thebenefit of transmitting at the correct polarization for each user.

[0236] Second, the trial results clearly demonstrated high correlationbetween the received signal levels for the two signals (representing theup- and down-link transmissions). Calculated correlation coefficients of90% and above resulted for the long-term raw data (non-smoothed). Thisindicates that polarization matching exists, for all channels tested inthe trials.

[0237] Overall, these results indicate the existence of the polarizationmatching, and sets the basis for its implementation in wireless mobilecommunication systems.

[0238] Based on the XPD and polarization matching trial resultssummarized above, it is expected that the following power values (withrespect to a vertical polarization transmitter site) may result fromapplying the polarization matching algorithms in the down- (forward-)link of cellular/PCS site equipment: Forward-link gain of 3-5 dB(average), and over 9 dB recovery from “coverage holes” (i.e.,−polarization mismatch). This stems from an average polarizationmismatch of 7 dB (and maximum of 9 dB) that is estimated to be thesituation of 95% of the users in a typical cellular sector, with someimplementation losses. This improvement is decoupled from fading, whichexists in a polarization mismatched system, and will continue to existin a polarization matched system. The fading observed all along theTrial runs followed its well-known behavior, including dependence on theradial vehicle speed, and independence between the two-frequencyseparated signals.

[0239] The trials provide conclusive results of the existence ofpolarization matching between two transmissions over the same channeland separated in frequency by at least as much as is common for FDMCellular/PCS wireless systems. Other systems, e.g. iDEN, TETRA, etc. mayalso benefit from application of the polarization matching technology.Additionally, the results demonstrate that the XPD is at least 10 dBover LOS and NLOS rural and sub-urban channels, thus justifying theimplementation of polarization matching algorithms withdual-polarization transmitters in various wireless mobile communicationsystems, such as in Cellular/PCS base-station equipment.

[0240] In cellular systems, the data required to detect and determinethe received signal polarization may be embedded within the Rx MODEMprocessing, as illustrated above. To use this data requires an interfacewith the Rx MODEM, on per-user basis. Because this interface is notalways available, another embodiment enables estimation of the receivedpolarization for TDMA system without an interface with the Rx MODEM. Theembodiment may use a timing signal, which signifies the timing of theserviced user. This timing signal may be used for the Transmitpolarization matching operation.

[0241]FIG. 13 illustrates a received vector representing RF signalpolarized in direction θ. It may be assumed that the RF signal level isnormalized and equal to 1. If the horizontal component is β, then thevertical component is {square root}{square root over (1−β²)}. Theembodiment estimates the long-term (averaged over many TDMA time slots)polarization direction (i.e., a long-term estimate of β, includingsign).

[0242] The output from each antenna may be transferred to a LNA and RFfilter. Then, input to a Σ−Δ RF combiner that will produce dual outputsthat are proportional to the sum of the two inputs and to the differencebetween them. Thus, considering the signal amplitude at each port, andassuming that the overall system gain is still unity, the outputs are:

Σ=β+{square root}{square root over (1−β²)}

Δ=β−{square root}{square root over (1−β²)}

[0243]FIG. 14 illustrates a graph of the absolute value (i.e., anenvelope detector) for each output. The graph shows the dependence ofABS (Σ) and ABS (Δ) versus β.

[0244]FIG. 15 illustrates the log ratio Σ/Δ that provides informationabout β, such as the sign of β. However, it is non-linear in β and,moreover, it does not provide one-to-one dependency.

[0245] In FIG. 16, the logarithmic representation of Σ/Δ providesinformation about the sign of the polarization vector (quadrantselection); however, it is not one-to-one dependent on β, and it ishighly non-linear. Thus, the absolute value (i.e., envelope detection)of the original signals is taken. The measurement is linearly dependenton β, but loses the sign (since the envelope of the signal is measured).When combining the results from the Σ/Δ measurement, an estimate for β,and quadrant identification can be found. FIG. 16 is a block diagram isan embodiment in which β is estimated.

[0246] It should be noted that β changes slowly with time. Thus, it ispossible to implement the same principle utilizing only one receiver andswitching between signals, for example, with two SP4T switches. In thisembodiment, the processor (i.e., digital signal processor) processes theoutput of the output switch, as shown in FIG. 17.

[0247] For forward links, polarization matching techniques are developedin embodiments of this invention that make BSs capable of transmittingsignals that match the polarization state of a mobile station duringBS-to-MS transmission.

[0248] While the invention has been described with reference to thecertain illustrated embodiments, the words which have been used hereinare words of description, rather than words of limitation. Changes maybe made, within the purview of the appended claims, without departingfrom the scope and spirit of the invention in its aspects. Although theinvention has been described herein with reference to particularstructures, acts, and materials, the invention is not to be limited tothe particulars disclosed, but rather extends to all equivalentstructures, acts, and materials, such as are within the scope of theappended claims.

What is claimed:
 1. A method of controlling a plurality of beam patternsradiated by a base station in a wireless communication system,comprising: receiving at least one signal from a mobile station at thebase station; determining estimated attributes of said at least onesignal received by the base station; calculating smoothed versions ofsaid estimated attributes according to a predetermined set of criteria;generating a set of weighted signal parameters according to saidsmoothed versions to describe a polarization state of said at least onesignal; and applying said weighted signal parameters to modify a signaltransmitted by said base station such that said transmitted signalsubstantially matches said polarization state of said at least onesignal.
 2. The method of claim 1, further comprising setting a defaultpolarization state of transmit elements in the base station.
 3. Themethod of claim 2, wherein said default polarization state is vertical.4. The method of claim 1, wherein the base station is configured for aTDMA wireless communication system and said weighted signal parametersare applied directly to said transmitted signal for each mobile station.5. The method of claim 1, wherein the base station is configured forCDMA wireless communication system and said weighted signal parametersare applied digitally at a baseband frequency of said CDMA wirelesscommunication system.
 6. The method of claim 1, wherein said estimatedattributes include at least amplitude and phase information of said atleast one signal.
 7. The method of claim 1, wherein said determining isperformed by a microprocessor.
 8. An apparatus, in a wirelesscommunication system, for controlling a plurality of beam patternsradiated by a base station to match the polarization of beam patternsradiated by at least one mobile station, comprising: an attributeestimation unit configured to process at least one signal received bythe base station; a data smoothing mechanism coupled to said attributeestimation unit, said data smoothing mechanism configured to calculatesmoothed versions of said estimated attributes in accordance with apredetermined set of criteria; and a normalized weight generator togenerate weighted signal parameters according to the smoothed versions,the weighted signal parameters being applied to said at least one signalto achieve polarization matching.
 9. The apparatus according to claim 8,wherein said attribute estimation unit produces a set of estimatedattributes, including at least an amplitude and phase value of said atleast one signal.
 10. The apparatus according to claim 8, furthercomprising: a quadrant detector configured to determine a sign of saidphase value; and a filtering unit configured to filter out short termvariations in said at least one signal based on said estimatedattributes, wherein said normalized weight generator is coupled to saidquadrant detector and said filtering unit.